In a conventional wireless signal processing circuit, discrete components are used for respective functional blocks (such as an amplifier for signal amplification, a mixer for signal frequency conversion, a filter for passing only a signal of a desired bandwidth). However, with the recent improvements in semiconductor technology, a plurality of functional blocks configuring a wireless signal processing circuit have been able to be incorporated in one semiconductor chip. A wireless signal processing circuit with its components incorporated in one or more semiconductor chips converts a radio-frequency signal received from an antenna to a signal of a lower frequency band with high quality (with less noise or suppressing signals in bands other than a desired band).
To achieve a wireless signal processing circuit at low cost, more functional blocks configuring the wireless signal processing circuit have to be incorporated in one semiconductor chip. One of the problems to be solved to achieve this object is that a filter circuit that suppresses signals in bands other than a desired band cannot be incorporated in a semiconductor chip. In general, an SAW (Surface Acoustic Wave) filter, a dielectric filter, or the like is used as the filter circuit. The signals in bands other than a desired band are suppressed by using such a filter. However, such an SAW filter or dielectric filter cannot be incorporated in a semiconductor chip.
In general, the wireless signal processing circuit formed of discrete components is formed in a configuration called superheterodyne method, which requires an SAW filter or dielectric filter (regarding superheterodyne, refer to, for example, “The Design of CMOS Radio-Frequency Integrated Circuits” by Thomas H. Lee, CAMBRIDGE (Non-Patent Document 1)). However, such an SAW filter or dielectric filter cannot be incorporated in a semiconductor chip. Therefore, if a wireless signal processing circuit fabricated through a semiconductor process is configured according to the superheterodyne method, an SAW filter or dielectric filter has to be externally provided outside a semiconductor chip. This increases the number of components and implementation area.
To get around this problem, a scheme for a wireless signal processing circuit without requiring an SAW filter or dielectric filter by using an advantage of a semiconductor circuit (although the absolute values of component constants are varied among semiconductor chips, relative values of the component constants in one semiconductor chip coincide with one another with high accuracy) has been proposed. This scheme includes a zero-IF scheme, a low-IF scheme, or a wideband IF scheme. In any of these schemes, an external SAW filter or dielectric filter is not required, and signals in bands other than the desired band are suppressed by a filter that can be incorporated in a semiconductor (there may be a case where part of the filter may have to be externally provided depending on a wireless scheme or due to a system requirement).
Basic principles of the zero-IF scheme, the low-IF scheme, the wideband IF scheme, and others are described in, for example, “DIRECT CONVERSION RECEIVERS IN WIDE-BAND SYSTEMS” by Aarno Parssinen, Kluwer Academic Publishers (Non-Patent Document 2). In the zero-IF scheme, the low-IF scheme, and the wideband IF scheme, there is a common characteristic in the configuration of a mixer circuit for signal frequency conversion. This mixer is referred to as a quadrature mixer, and its general circuit diagram is shown in FIG. 2.
In the quadrature mixer circuit of FIG. 2, 10 denotes a local signal oscillator; 20 denotes a 90-degree phase-shift circuit, 30 to 50 denote bias circuits; VCC denotes a battery; C3 and C4 denote capacitors; R1, R2, RI1, RI2, RI5 to RI7, RQ1, RQ2, RQ5 to RQ7 denote resistors; QI1 to QI6 and QQ1 to QQ6 denote transistors; IEI1, IEI2, IEQ1, and IEQ2 denote current sources; RFin1 and RFin2 denote input terminals; Iout1 and Iout2 denote I output terminals; and Qout1 and Qout2 denote Q output terminals.
In FIG. 2,
the current of the current source IEI1=the current of the current source IEI2=the current of the current source IEQ1=the current of the current source IEQ2;
the transistors QI1, QI2, QQ1, and QQ2 are equal to one another in size, shape, and characteristic;
the transistors QI3 to QI6 and QQ3 to QQ6 are equal to one another in size, shape, and characteristic;
the resistance value of RI1=the resistance value of RI2=the resistance value of RQ1=the resistance value of RQ2=RL;
the resistance value of RI6=the resistance value of RI7=the resistance value of RQ6=the resistance value of RQ7;
the resistance value of RI5=the resistance value of RQ5=RE;
the resistance value of R1=the resistance value of R2; and
the capacitance value of C3=the capacitance value of C4.
It is assumed that the bias circuits 30 to 50 provide a direct-current (DC) bias voltage to the transistors QI1 to QI6 and QQ1 to QQ6 of FIG. 2 so that the transistors QI1 to QI6 and QQ1 to QQ6 operate appropriately. That is, it is assumed that:
the base voltage of the transistor QI1=the base voltage of the transistor QI2=the base voltage of the transistor QQ1=the base voltage of the transistor QQ2;
the base voltage of the transistor QI3=the base voltage of the transistor QI4=the base voltage of the transistor QI5=the base voltage of the transistor QI6=the base voltage of the transistor QQ3=the base voltage of the transistor QQ4=the base voltage of the transistor QQ5=the base voltage of the transistor QQ6;
the base voltage of the transistor QI1>the emitter voltage of the transistor QI1;
the collector voltage of the transistor QI1>the emitter voltage of the transistor QI1;
the base voltage of the transistor QI2>the emitter voltage of the transistor QI2;
the collector voltage of the transistor QI2>the emitter voltage of the transistor QI2;
the collector voltages of the transistors QI3 and QI5>the emitter voltages of the transistors QI3 and QI5;
the collector voltages of the transistors QI4 and QI6>the emitter voltages of the transistors QI4 and QI6;
the base voltage of the transistor QQ1>the emitter voltage of the transistor QQ1;
the collector voltage of the transistors QQ1>the emitter voltage of the transistor QQ1;
the base voltage of the transistor QQ2>the emitter voltage of the transistors QQ2;
the collector voltage of the transistor QQ2>the emitter voltage of the transistor QQ2;
the collector voltages of the transistors QQ3 and QQ5>the emitter voltages of the transistors QQ3 and QQ5; and
the collector voltages of the transistors QQ4 and QQ6>the emitter voltages of the transistors QQ4 and QQ6.
Direct-current components of an RF signal inputted as a differential signal from the input terminal RFin1 are cut by the capacitor C3, and the RF signal is then branched and inputted to the bases of the transistors QI2 and QQ2. Also, direct-current components of an RF signal inputted as a differential signal from the input terminal RFin2 are cut by the capacitor C4, and the RF signal is then branched and inputted to the bases of the transistors QI1 and QQ1. The current sources IEI1 and IEI2 supply a bias current to the transistors QI1 and QI2, respectively, and the current sources IEQ1 and IEQ2 supply a bias current to the transistors QQ1 and QQ2, respectively.
The RF signal voltages inputted as differential signals from the input terminals RFin1 and RFin2 are converted to the RF signal currents through the voltage-current conversion by the transistors QI1, QI2, QQ1 and QQ2 and the resistors RI5 and RQ5.
When an RF signal voltage (alternating-current component) inputted as a differential signal from the input terminals RFin1 and RFin2 is taken as vRF and an RF signal current after voltage-current conversion (alternating-current component) is taken as iRF,
iRF
=the RF signal current of the collector current of the transistor QI1 (alternating-current component)-the RF signal current of the collector current of the transistor QI2 (alternating-current component)
=the RF signal current of the collector current of the transistor QQ1 (alternating-current component)-the RF signal current of the collector current of the transistor QQ2 (alternating-current component)
=vRF/RE
Local signals outputted from the local signal oscillator 10 are inputted to the bases of the transistors QI3 to QI6. Therefore, the transistors QI3 to QI6 operate as switching circuits that switch a current with the same phase as that of the local signal oscillator 10. In this case, the current to be switched is iRF described above. By switching iRF with the same phase as that of the local signal, a difference component between the collector voltage of the transistor QI3 or QI5 and the collector voltage of the transistor QI4 or QI6 (VIout1-VIout2 when a voltage of the output terminal Iout1 is taken as VIout1 and a voltage of the output terminal Iout2 is taken as VIout2) includes frequency components indicative of difference and sum of the signal frequency components of iRF and the signal frequency of the local signal oscillator 10.
The local signal is a differential signal. That is, the local signal inputted to the base of the transistor QI3 or QI5 is different in phase by 180 degrees from the local signal inputted to the base of the transistor QI4 or QI6. When the amplitude voltage of the local signal inputted to the base of the transistor QI3 or QI5 is larger than the amplitude voltage of the local signal inputted to the base of the transistor QI4 or QI6, the state between the base and the collector of the transistor QI3 or QI5 is in an ON state, and the state between the base and the collector of the transistor QI4 or QI6 is in an OFF state. When the amplitude voltage of the local signal inputted to the base of the transistor QI3 or QI5 is smaller than the amplitude voltage of the local signal inputted to the base of the transistor QI4 or QI6, the state between the base and the collector of the transistor QI3 or QI5 is in an OFF state, and the state between the base and the collector of the transistor QI4 or QI6 is in an ON state. In this manner, the switching operation of iRF is performed.
Filters (not shown) are provided at stages subsequent to Iout1 and Iout2, and only the desired frequency components are extracted from the frequency components included in VIout1-VIout2 and indicative of the difference and sum of the signal frequency components of iRF and the signal frequency of the local signal oscillator 10.
Local signals obtained by shifting the phase of the signal of the local signal oscillator 10 by 90 degrees at the 90-degree phase-shift circuit 20 are inputted to the bases of the transistors QQ3 to QQ6. Therefore, the transistors QQ3 to QQ6 operate as switching circuits that switch a current with the phase different by 90 degrees from the phase of the local signal oscillator 10. In this case, the current to be switched is iRF described above. By switching iRF with the phase different by 90 degrees from that of the local signal, a difference component between the collector voltage of the transistor QQ3 or QQ5 and the collector voltage of the transistor QQ4 or QQ6 (VQout1-VQout2 when a voltage of the output terminal Qout1 is taken as VQout1 and a voltage of the output terminal Qout2 is taken as VQout2) includes frequency components indicative of difference and sum of the signal frequency components of iRF and the signal frequency of the local signal oscillator 10.
As described above, since the local signal is a differential signal, the local signal inputted to the base of the transistor QQ3 or QQ5 is different in phase by 180 degrees from the local signal inputted to the base of the transistor QQ4 or QQ6. When the amplitude voltage of the local signal inputted to the base of the transistor QQ3 or QQ5 is larger than the amplitude voltage of the local signal inputted to the base of the transistor QQ4 or QQ6, the state between the base and the collector of the transistor QQ3 or QQ5 is in an ON state, and the state between the base and the collector of the transistor QQ4 or QQ6 is in an OFF state. When the amplitude voltage of the local signal inputted to the base of the transistor QQ3 or QQ5 is smaller than the amplitude voltage of the local signal inputted to the base of the transistor QQ4 or QQ6, the state between the base and the collector of the transistor QQ3 or QQ5 is in an OFF state, and the state between the base and the collector of the transistor QQ4 or QQ6 is in an ON state. In this manner, the switching operation on iRF is performed.
Filters (not shown) are provided at stages subsequent to Qout1 and Qout2, and only the desired frequency components are extracted from the frequency components included in VQout1-VQout2 and indicative of the difference and sum of the signal frequency component of iRF and the signal frequency of the local signal oscillator 10.
In a circuit portion for voltage-current conversion of the RF signal voltage to the RF signal current in the quadrature mixer circuit of FIG. 2, that is, in a portion configured of the transistors QI1, QI2, QQ1, and QQ2, the resistors RI5 and RQ5, and the current sources IEI1, IEI2, IEQ1 and IEQ2, a portion configured of the transistors QI1 and QI2, the resistor RI5, and the current sources IEI1 and IEI2 and a portion configured of the transistors QQ1 and QQ2, the resistor RQ5, and the current sources IEQ1 and IEQ2 perform the same voltage-current conversion operation. These portions performing the same operation are combined together so as to reduce current consumption.
An example of a low-current-consumption quadrature mixer circuit is shown in FIG. 3. In FIG. 3, portions performing operations similar to those of portions in FIG. 2 are provided with the same reference numerals and are not described here.
In the quadrature mixer circuit of FIG. 3, IE1 and IE2 denote current sources, and R5, RI3, RI4, RQ3, and RQ4 denote resistors. In FIG. 3, in addition to the bias conditions of FIG. 2, the following conditions are provided, that is:
the current of the current source IE1=the current of the current source IE2;
the emitter current of the transistor QI1+the emitter current of the transistor QQ1=the current of the current source IE1;
the emitter current of the transistor QI2+the emitter current of the transistor QQ2=the current of the current source IE2;
the emitter current of the transistor QI1=the emitter current of the transistor QQ1;
the emitter current of the transistor QI2=the emitter current of the transistor QQ2; and
the resistance value of RI3=the resistance value of RI4=the resistance value of RQ3=the resistance value of RQ4=RC.
In the quadrature mixer circuit of FIG. 3, the RF signal voltages inputted as differential signals from the input terminals RFin1 and RFin2 are converted to RF signal currents (alternating-current components) through voltage-current conversion by the transistors QI1, QI2, QQ1, and QQ2, and the resistor R5, and the RF signal currents are branched at the resistors RI3 and RI4 and the resistors RQ3 and RQ4 and then inputted to the transistors QI3 and QI4 and the transistors QQ3 and QQ4 of the current switching units. Therefore, this quadrature mixer circuit can operate with half of the current consumption of the quadrature mixer circuit of FIG. 2 (however, the current consumption may not always become exactly half depending on adjustment of the circuit at a design stage). At this time, the resistors RI3 and RI4 and the resistors RQ3 and RQ4 are inserted in the portions which switch the current, that is, the transistors QI3 to QI6 and the transistors QQ3 to QQ6, respectively, so that the local signals inputted to the respective bases of the transistors do not interfere with one another. The resistors RI3 and RI4 and the resistors RQ3 and RQ4 are described in Japanese patent application laid-open publication No. 2004-180281 (Patent Document 1).